Linear capacitance measurement and touchless switch

ABSTRACT

Capacitance measurement apparatus that enhances the sensitivity and accuracy of capacitive transducers, proximity sensors, and touchless switches. Each of two capacitors (C 1,  C 2 ) under measurement has one end connected to ground and is kept at substantially the same voltage potential by operational amplifier (A 1 ) or amplifiers (A 0,  A 1 ) using negative feedback. The apparatus is driven by a periodic e.g. sinusoidal signal source (G 1 ) or sources (G 1,  G 2 ) and includes a difference amplifier (A 2 ) operative to produce an electrical signal having a linear relationship with a specified arithmetic function of the capacitances of the two capacitors (C 1,  C 2 ). A touchless switch is implemented using the capacitance measurement apparatus. The touchless switch includes two sensor electrodes (E 1,  E 2 ) that correspond to the two capacitors (C 1,  C 2 ) under measurement and in one embodiment has a front surface in the form of a container.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part (CIP) application of priorU.S. patent application Ser. No. 11/202,486 filed Aug. 12, 2005 entitledLINEAR CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH. This applicationclaims benefit of U.S. Provisional Patent Application No. 60/690,486filed Jun. 15, 2005 entitled LINEAR CAPACITANCE MEASUREMENT ANDTOUCHLESS SWITCH, U.S. Provisional Patent Application No. 60/662,378filed Mar. 17, 2005 entitled CAPACITANCE MEASUREMENT AND TOUCHLESSSWITCH, U.S. Provisional Patent Application No. 60/619,697 filed Oct.19, 2004 entitled DIFFERENTIAL CAPACITANCE MEASUREMENT AND TOUCHLESSSWITCH, and U.S. Provisional Patent Application No. 60/601,610 filedAug. 16, 2004 entitled DIFFERENTIAL CAPACITANCE MEASUREMENT ANDTOUCHLESS SWITCH.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

Not applicable

BACKGROUND OF THE INVENTION

The present invention relates generally to capacitance measurementapparatus and techniques, and more specifically to proximity detectorssuch as touchless switches that employ capacitance measurementtechniques.

In recent years, there has been an increasing need for improvedtechniques of operating publicly accessible facilities and equipmentwithout requiring a user to make physical contact with a surface of amanual activation device such as a touch switch. Such facilities andequipment include elevators, vending machines, security access panels,information terminals, etc. By not requiring a user to physically toucha switch that may have been touched and contaminated by others who hadpreviously used the facilities or equipment, the spread of germs anddiseases may be significantly reduced.

For example, a user typically operates a public facility such as anelevator by physically touching one or more switches, which may havebeen previously touched by a substantial number of individuals. Some ofthese individuals may have come from environments where they may havebeen exposed to contaminants such as potentially harmful or contagioustoxins or pathogenic disease organisms. When such individuals makephysical contact with one or more of the switches required to operate anelevator, there is a risk that the individuals may deposit contaminantsonto the surface of the switches, where they may remain viable for anextended period of time. These contaminants may be later transferredfrom the switches to subsequent elevator users who physically touch theswitches, thereby potentially causing the subsequent users to becomeafflicted with diseases or other serious medical conditions.

During outbreaks of the severe acute respiratory syndrome (SARS) inAsia, many members of the public were afraid to use any publicfacilities that required them to touch a manual activation device suchas a touch switch. To mitigate the fears of the public, programs wereinstituted for periodically cleaning and disinfecting the surfaces ofthese devices. Such programs are typically ineffective, because nomatter how well these activation devices are cleaned and disinfected,they may become contaminated once again by a subsequent user. As aresult, the risk of transferring potentially harmful contaminants frommanual activation devices such as touch switches to subsequent users ofpublicly accessible facilities and equipment continues unabated.Capacitance-based proximity detectors have been employed to implementactivation devices that do not require a user to physically touch asurface of the device. Such proximity detectors operate according to theprinciple that an electric field and a capacitance are generated betweentwo conductive objects that have different voltage potentials and arephysically separated from one another. The capacitance between the twoconductive objects generally increases as the surface areas of theobjects increase, or as the distance between the objects decreases.

Conventional capacitance-based proximity detectors have drawbacks,however, when they are used to implement a touchless switch. Forexample, it is generally difficult to adjust the sensitivity of acapacitance-based proximity detector to assure that a touchless switchemploying such a proximity detector can be reliably activated by a broadrange of users, and that the switch is not susceptible to noise and/orenvironmental changes. This is due to the relatively small equivalentcapacitance that the capacitance-based proximity detector is required tomeasure when implementing a touchless switch.

Specifically, when a human body is very near or proximate to a sensorelectrode of a capacitance-based proximity detector, the proximitydetector effectively measures the equivalent capacitance of two seriescapacitors, assuming that the stray capacitance between the capacitancesensing circuitry and circuit ground is ignored. One of the seriescapacitors is formed between the sensor electrode and the human body,and the other capacitor is formed between the human body and earthground. The amount of capacitance between the sensor electrode and thehuman body depends primarily on the distance between them, and to alesser extent on the size and characteristics of the human body. Forexample, when the human body is not very near the sensor electrode, theamount of capacitance between the sensor electrode and the human body issignificantly smaller than the amount of capacitance between the humanbody and ground. Accordingly, a touchless switch implemented using acapacitance-based proximity detector must measure an equivalentcapacitance that is significantly smaller than the capacitance typicallymeasured by a conventional touch switch.

FIG. 1 depicts a touchless switch implemented using a capacitance-basedproximity detector 100 including a sensor electrode 112, capacitancesensing circuitry 114, and the equivalent capacitances of the capacitorsformed between a human finger and the sensor electrode 112 (C_(A)), therest of the human body and the sensor electrode 112 (C_(B)), the humanbody and ground (C_(C)), and the capacitance sensing circuitry 114 andground (C_(D)), which in this analysis can be ignored. When the humanfinger is proximate to the sensor electrode 112, the capacitance betweenthe human body and the sensor electrode 112 can be taken as the sum ofthe capacitance C_(A) between the finger and the sensor electrode 112,and the capacitance C_(B) between the rest of the human body and thesensor electrode 112. If the human finger is not very near the sensorelectrode 112, then any changes in the capacitance C_(A) between thefinger and the sensor electrode 112 are typically very small. As aresult, any extraneous common-mode disturbances resulting fromelectrical noise or interference, changes in the characteristics of theenvironment, changes in the capacitance C_(C) between the human body andground, and/or changes in the capacitance C_(B) between the rest of thehuman body and the sensor electrode 112 due to changes in the distancebetween the rest of the human body and the sensor electrode 112, changesin the size or characteristics of the human body, etc., may be equal toor greater than the corresponding changes in the capacitance C_(A)between the human finger and the sensor electrode 112. Accordingly, ifthe sensitivity of the capacitance-based proximity detector 100 isadjusted to be highly sensitive, then the proximity detector 100 may beactuated unintentionally, due to the various extraneous common-modedisturbances listed above. However, if the capacitance-based proximitydetector 100 has reduced sensitivity, then the proximity detector 100may be inoperable due to the inability to detect the small amount ofcapacitance between the finger of a user and the sensor electrode 112 ata reasonable distance.

A touch switch implemented using the capacitance-based proximitydetector 100 generally fares much better than a touchless switch becausewhen a human finger touches the surface of a touch switch, the area ofcontact is typically much larger than just the area of a fingertip.Further, the distance between a finger and a sensor electrode of thetouch switch is typically much smaller than the corresponding distancebetween a finger and the sensor electrode 112 of the touchless switch,even if the sensor electrode of the touch switch is disposed behind aninsulating surface. The changes in capacitance between a human fingerand the sensor electrode of a touch switch are therefore much largerthan the corresponding changes in capacitance between a human finger andthe sensor electrode 112 of a touchless switch. Accordingly, theproblems described above relating to the detection of changes in thecapacitance C_(A) between a human finger and the sensor electrode 112 ofthe touchless switch, e.g., the changes in the capacitance C_(B) orC_(C) due to different users, are relatively insignificant in a touchswitch.

One way of avoiding the problems described above relating to extraneouscommon-mode disturbances in a touchless switch is to employ knowndifferential signal measurement techniques. Such differential signalmeasurement techniques can be used in touchless switches that includetwo sensor electrodes arranged so that the switch is actuated when thecapacitance between a human finger and one of the sensor electrodesexceeds a preset threshold level relative to a second capacitancebetween the finger and the other sensor electrode. By directly comparingthese first and second capacitances in a differential measurement todetermine whether to actuate the touchless switch, extraneouscommon-mode disturbances that can adversely affect the measurement canbe effectively canceled out.

U.S. Pat. No. 6,310,611 filed Oct. 30, 2001 entitled DIFFERENTIAL TOUCHSENSOR AND CONTROL CIRCUIT THEREFORE (the '611 patent) discloses a touchsensor that employs a differential signal measurement technique. Asdisclosed in the '611 patent, the touch sensor includes a first sensorelectrode, a second sensor electrode positioned proximate to the firstelectrode, a differential circuit connected to the first and secondelectrodes, and a pulse or other signal source configured to generate anelectric field between the first and second electrodes. Although thetouch sensor of the '611 patent is configured to perform a differentialmeasurement, the touch sensor does not operate by measuring capacitance.Instead, the touch sensor measures changes in the voltage differencebetween the two sensor electrodes caused by the introduction of anobject affecting the electric field around the two electrodes. The touchsensor employs a differential circuit to provide an output signal thatis responsive to this difference in voltage between the two electrodes.

The touch sensor disclosed in the '611 patent has drawbacks, however,when used to implement a touchless switch. For example, theabove-described changes in the voltage difference between the two sensorelectrodes of the touch sensor resulting from the introduction of anobject are caused by the interaction of the electric fields associatedwith the sensor electrodes and the object. This interaction of electricfields is relatively complex because the two sensor electrodes and theobject are at different voltage potentials, and there is no preciserelationship governing the voltage difference between the sensorelectrodes and the proximity of the object to the sensor electrodes.Furthermore, the methods disclosed in the '611 patent to measure thevoltage difference between the sensor electrodes are only effective ifthe voltage difference is significant enough as in the case of a touchswitch. Therefore, the approach disclosed in the '611 patent is notprecise or sensitive enough to be used in a touchless switch. U.S. Pat.No. 6,456,477 filed Sep. 24, 2002 entitled LINEAR CAPACITANCE DETECTIONCIRCUIT (the '477 patent) discloses capacitance detection circuitry thatemploys a differential signal measurement technique. As disclosed in the'477 patent, the linear capacitance detection circuitry includes acircuit that measures a difference in capacitance between a firstcapacitor and a second capacitor by driving the two capacitors withpulses. The capacitance detection circuitry further includes anoperational amplifier with negative feedback configured to maintain thetwo capacitors at substantially equal voltage potentials. As a result,there is a linear relationship between an electrical signal produced bythe operational amplifier and the ratio of the capacitances of the twocapacitors. The approach disclosed in the '477 patent also hasdrawbacks, however, in that it requires pulse signals, which canintroduce transient noises and instability to the operational amplifierand can adversely affect the accuracy of the operational amplifieroutput. Although low pass filters and a feedback capacitor may beemployed at the inputs of the operational amplifier to mitigate theeffects of transient noises and instability, the addition of suchcomponents adversely affects the accuracy and sensitivity of thecapacitance detection circuitry.

It would therefore be desirable to have a capacitance measurementapparatus and technique, and a proximity detector such as a touchlessswitch employing a capacitance measurement technique, that avoid thedrawbacks of the above-described approaches.

BRIEF SUMMARY OF THE INVENTION

In accordance with the present invention, a capacitance measurementapparatus and technique are provided that can be employed to enhance thesensitivity and accuracy of many different types of capacitivetransducers, proximity sensors, and touchless switches. The presentlydisclosed capacitance measurement apparatus directly and accuratelyproduces a linear response to changes in each of the ratios of thecapacitance of a capacitor/capacitive transducer to the capacitance ofone or more other different capacitors/capacitive transducers withadjustable offset, while maintaining all of the capacitors/capacitivetransducers at substantially identical voltage potentials at all times.The presently disclosed capacitance measurement apparatus also producesa linear response to changes in each of the differences between thecapacitance of a capacitor/capacitive transducer multiplied by a firstconstant factor and the capacitance of one or more other differentcapacitors/capacitive transducers each being multiplied by a respectivesecond constant factor, while maintaining all of thecapacitors/capacitive transducers at substantially identical voltagepotentials at all times.

Additionally, the presently disclosed capacitance measurement apparatusdirectly and accurately produces a linear response to changes in thecapacitance, or changes in the reciprocal of the capacitance, of acapacitor/capacitive transducer with adjustable offset, withoutrequiring special calibration or adjustment over a wide range ofcapacitance values. The presently disclosed capacitance measurementapparatus also provides a simple way of measuring the capacitance, orthe reciprocal of the capacitance, of a large number ofcapacitors/capacitive transducers, or comparing the capacitance of alarge number of capacitors/capacitive transducers with the capacitanceof a large number of sets of capacitors/capacitive transducers.

The presently disclosed capacitance measurement apparatus employs aplurality of operational amplifiers for maintaining the voltagepotentials of multiple capacitors/capacitive transducers undergoingcomparison or measurement at substantially the same voltage potential atall times. Because the multiple capacitors/capacitive transducers aremaintained at substantially the same voltage potential, there isessentially no capacitance between them. For this reason, thecapacitance measurement apparatus can be employed to measure smallchanges in the capacitances of the multiple capacitors/capacitivetransducers without having adjacent capacitors/capacitive transducersaffect the capacitance measurement, even when the capacitors/capacitivetransducers are positioned relatively close to one another. In oneembodiment, the capacitance measurement apparatus includes a firstoperational amplifier Al, and a second operational amplifier A2configured as a difference amplifier. A difference amplifier, as theterm is used herein, refers to a circuit or device that amplifies adifference between two input signals and includes different types ofdifferential DC amplifiers such as instrumentation amplifiers, etc. Eachof two capacitors C1 and C2 undergoing comparison or measurement has oneend connected to circuit ground, and another end connected to one of thedifferential inputs of operational amplifier A1. Capacitor C1 isconnected to the inverting input of operational amplifier A1, andcapacitor C2 is connected to the non-inverting input of operationalamplifier A1. Both capacitors C1 and C2 are driven by the output of aperiodic varying voltage source such as a sinusoidal voltage sourcethrough respective resistors connected to corresponding inputs ofoperational amplifier A1. A feedback resistor is connected between theoutput of operational amplifier Al and its inverting input. Due to thehigh open loop gain of operational amplifier A1, capacitors C1 and C2are maintained at substantially the same voltage potential at all times.There is a linear relationship between the magnitude of the currentflowing through the feedback resistor and the ratio of the capacitanceof capacitor C1 to the capacitance of capacitor C2. Further, the currentflowing through the feedback resistor is in-phase or out-of-phase withthe currents flowing through the resistors connected to the periodicvarying voltage source, depending on whether the ratio is less than orgreater than a specified value. The phase and magnitude of the currentflowing through the feedback resistor can be measured by differenceamplifier A2, having one of its differential inputs connected to theoutput of operational amplifier Al and another differential inputconnected to one of the differential inputs of operational amplifier A1.

In a second embodiment, the capacitance measurement apparatus includes afirst operational amplifier A1, and a second operational amplifier A2configured as a difference amplifier. Each of two capacitors C1 and C2undergoing comparison or measurement has one end connected to circuitground, and another end connected to one of the differential inputs ofoperational amplifier A1. Capacitor C1 is connected to the invertinginput of operational amplifier A1, and capacitor C2 is connected to thenon-inverting input of operational amplifier A1. The non-inverting inputof operational amplifier Al is driven directly by the output of a firstperiodic varying current source such as a sinusoidal current source,while the inverting input of operational amplifier Al is driven directlyby a second periodic varying current source whose output is K (aconstant) times that of the first periodic varying current source. Afeedback resistor is connected between the output of operationalamplifier A1 and its inverting input. Due to the high open loop gain ofoperational amplifier A1, capacitors C1 and C2 are maintained atsubstantially the same voltage potential at all times. There is a linearrelationship between the magnitude of the current flowing through thefeedback resistor and the ratio of the capacitance of capacitor C1 tothe capacitance of capacitor C2. Further, the current flowing throughthe feedback resistor is in-phase or out-of-phase with the outputs ofthe periodic varying current sources, depending on whether the ratio isless than or greater than the value of K. The phase and magnitude of thecurrent flowing through the feedback resistor can be measured bydifference amplifier A2, having one of its differential inputs connectedto the output of operational amplifier A1 and another differential inputconnected to one of the differential inputs of operational amplifier A1.

In a third embodiment, the capacitance measurement apparatus includesfirst and second operational amplifiers A0 and A1, and a thirdoperational amplifier A2 configured as a difference amplifier. Each oftwo capacitors C1 and C2 undergoing comparison or measurement, havingcapacitances of c1 and c2, respectively, has one end connected tocircuit ground, and another end connected to the inverting input ofoperational amplifier A0 or A1. Capacitor C1 is connected to theinverting input of operational amplifier A1, and capacitor C2 isconnected to the inverting input of operational amplifier A0. Thenon-inverting inputs of operational amplifiers A0 and A1 are both drivendirectly by a periodic varying voltage source such as a sinusoidalvoltage source. A first feedback resistor R1 having a resistance of r1is connected between the output of operational amplifier A1 and itsinverting input. A second feedback resistor R2 having a resistance of r2is connected between the output of operational amplifier A0 and itsinverting input. Due to the high open loop gain of operationalamplifiers A0 and A1, the two capacitors C1 and C2 are maintained atsubstantially the same voltage potential as the periodic varying voltagesource at all times. The output of operational amplifier A1 is connectedto the non-inverting input of difference amplifier A2, and the output ofoperational amplifier A0 is connected to the inverting input ofdifference amplifier A2. The output of difference amplifier A2 isproportional to (r1*c1-r2*c2), and is in-phase or out-of-phase with thecurrents flowing through resistors R1 and R2, depending on whether(r1*c1-r2*c2) is greater than or less than zero.

Each embodiment of the presently disclosed capacitance measurementapparatus can be configured for comparing the capacitance of acapacitor/capacitive transducer to the capacitance of a plurality ofother different capacitors/capacitive transducers, while maintaining allof the capacitors/capacitive transducers at substantially the samevoltage potential. Additionally, by switching the respectivecapacitors/capacitive transducers in and out for subsequent comparisonor measurement, each embodiment of the capacitance measurement apparatuscan sequentially measure the capacitance, or the reciprocal of thecapacitance, of multiple capacitors/capacitive transducers, or comparethe capacitance of multiple capacitors/capacitive transducers to thecapacitance of multiple sets of capacitors/capacitive transducers.

Touchless switches and proximity sensors are also provided that employembodiments of the presently disclosed capacitance measurementapparatus. The touchless switches are configured to be actuated by ahuman finger or a finger-like object, requiring the finger orfinger-like object to reach a specified boundary before actuating theswitch. The touchless switches have reduced susceptibility to unintendedactuations, and reduced sensitivity to changes in environmental factorssuch as temperature, humidity, etc., and to electrical noise, whilehaving a simple and rugged construction. The touchless switches can beused in hygiene-sensitive applications, industrial control panels, and awide variety of facilities and equipment accessible to the generalpublic, including but not limited to elevators, vending machines,security access panels, information terminals, etc.

In one embodiment, the touchless switch includes a front surface, andtwo adjacent sensor electrodes maintained at substantially the samevoltage potential disposed on or behind the front surface of the switch.As a result, there is substantially no capacitance between the twosensor electrodes, and therefore the sensor electrodes operateessentially independent of one another. One of the sensor electrodes isa center electrode, and the other sensor electrode is an outerelectrode. The center electrode is spaced from and at least partlysurrounded by the outer electrode. When the tip of a human finger orfinger-like object is near or proximate to the center electrode, thepresence of the finger or finger-like object can be detected using anembodiment of the capacitance measurement apparatus disclosed herein.The capacitance measurement apparatus can be employed to measure theratio of the capacitance of the two sensor electrodes with respect tothe finger or finger-like object, or the difference between thecapacitance of one sensor electrode with respect to the finger orfinger-like object multiplied by a first constant factor, and thecapacitance of the other sensor electrode with respect to the finger orfinger-like object multiplied by a second constant factor, therebysubstantially canceling out extraneous common-mode disturbances such asthe capacitance between the rest of the human body and the sensorelectrodes, the capacitance between the human body and earth ground,environmental changes, electrical noise, etc., which tend to affect bothsensor electrodes equally due to their close proximity to one another.The capacitance ratio and difference measurements performed by thecapacitance measurement apparatus are facilitated by the fixedgeometrical shape, size, and relative positions of the two sensorelectrodes. The outer electrode may be placed in front of the centerelectrode so that initially, when the finger or finger-like object movestoward the center electrode, the capacitance ratio or differencemeasurement is less than a preset threshold. As the finger orfinger-like object moves closer to the center electrode, the capacitanceratio or difference measurement eventually exceeds the preset threshold,thereby actuating the switch. The touchless switch may also include aguard electrode surrounding the back and sides of the two sensorelectrodes. The guard electrode and the sensor electrodes are maintainedat substantially the same voltage potential so that each sensorelectrode forms a capacitor only with objects disposed in front of it.Leads extending from the two sensor electrodes to the capacitancemeasurement apparatus may also be guarded using a twin-axial cable ortwo coaxial cables, in which the outer conductors of the cables areemployed as guard shields and maintained at substantially the samevoltage potential as the inside cable conductors connected to the sensorelectrodes. The front surface of the touchless switch may take the formof the surface of a container, in which the brim of the containersurface defines an imaginary boundary plane that the finger orfinger-like object must reach to actuate the switch.

The presently disclosed capacitance measurement apparatus may beemployed to detect the proximity of conductive objects larger than ahuman finger such as the palm of a human hand. The presently disclosedcapacitance measurement apparatus may also be employed to detect thelocation, position, and/or movement of a conductive object such as ahuman appendage within a specified area.

Other features, functions, and aspects of the invention will be evidentfrom the Detailed Description of the Invention that follows.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

The invention will be more fully understood with reference to thefollowing Detailed Description of the Invention in conjunction with thedrawings of which:

FIG. 1 illustrates various equivalent capacitances formed between ahuman body, earth ground, and a sensor electrode coupled to capacitancesensing circuitry;

FIG. 2 a is a schematic diagram of first capacitance measurementcircuitry according to the present invention;

FIG. 2 b is a schematic diagram of circuitry employing the firstcapacitance measurement circuitry of FIG. 2 a, for producing a linearresponse to changes in each of the ratios of the capacitance of acapacitor to the capacitance of one or more other different capacitors;

FIG. 3 a is a schematic diagram of second capacitance measurementcircuitry according to the present invention;

FIG. 3 b is a schematic diagram of circuitry employing the secondcapacitance measurement circuitry of FIG. 3 a, for producing a linearresponse to changes in each of the ratios of the capacitance of acapacitor to the capacitance of one or more other different capacitors;

FIG. 4 a is a schematic diagram of third capacitance measurementcircuitry according to the present invention;

FIG. 4 b is a schematic diagram of circuitry employing the thirdcapacitance measurement circuitry of FIG. 4 a, for producing a linearresponse to changes in each of the differences between the capacitanceof a capacitor multiplied by a first constant factor and the capacitanceof one or more other different capacitors, after each is multiplied by arespective second constant factor;

FIGS. 5 a-5 d are perspective views of illustrative shapes of a frontsurface of a touchless switch according to the present invention;

FIGS. 6 a-6 d are cross-sectional views of illustrative arrangements andrelative positions of two sensor electrodes and the front surfaces ofthe touchless switches of FIGS. 5 a-5 d, respectively;

FIG. 7 is a cross-sectional view of an illustrative arrangement andrelative positions of two sensor electrodes, a front surface, and aguard electrode of a touchless switch according to the presentinvention;

FIG. 8 a is a diagram of a touchless switch employing the firstcapacitance measurement circuitry of FIG. 2 a;

FIG. 8 b is a diagram of a set of touchless switches employing the firstcapacitance measurement circuitry of FIG. 2 a;

FIG. 9 a is a diagram of a touchless switch employing the secondcapacitance measurement circuitry of FIG. 3 a;

FIG. 9 b is a diagram of a set of touchless switches employing thesecond capacitance measurement circuitry of FIG. 3 a;

FIG. 10 a is a diagram of a touchless switch employing the thirdcapacitance measurement circuitry of FIG. 4 a; and

FIG. 10 b is a diagram of a set of touchless switches employing thethird capacitance measurement circuitry of FIG. 4 a.

DETAILED DESCRIPTION OF THE INVENTION

The entire disclosures of U.S. patent application Ser. No. 11/202,486filed Aug. 12, 2005 entitled LINEAR CAPACITANCE MEASUREMENT ANDTOUCHLESS SWITCH, U.S. Provisional Patent Application No. 60/690,486filed Jun. 15, 2005 entitled LINEAR CAPACITANCE MEASUREMENT ANDTOUCHLESS SWITCH, U.S. Provisional Patent Application No. 60/662,378filed Mar. 17, 2005 entitled CAPACITANCE MEASUREMENT AND TOUCHLESSSWITCH, U.S. Provisional Patent Application No. 60/619,697 filed Oct.19, 2004 entitled DIFFERENTIAL CAPACITANCE MEASUREMENT AND TOUCHLESSSWITCH, and U.S. Provisional Patent Application No. 60/601,610 filedAug. 16, 2004 entitled DIFFERENTIAL CAPACITANCE MEASUREMENT ANDTOUCHLESS SWITCH, are incorporated herein by reference.

A capacitance measurement apparatus and technique are disclosed that canbe employed to enhance the sensitivity and accuracy of many differenttypes of capacitive transducers, proximity sensors, and touchlessswitches. FIG. 2 a depicts a first illustrative embodiment ofcapacitance measurement circuitry 200 a, in accordance with the presentinvention. In the illustrated embodiment, the capacitance measurementcircuitry 200 a includes a periodic varying voltage source G1, a firstoperational amplifier A1, and a second operational amplifier A2configured as a difference amplifier. Each of two capacitors/capacitivetransducers C1 and C2 undergoing comparison, having capacitances of c1and c2, respectively, has one end connected to circuit ground, andanother end connected to one of the differential inputs of operationalamplifier A1. Capacitor C1 is connected to the inverting input ofoperational amplifier A1 at node 101, and capacitor C2 is connected tothe non-inverting input of operational amplifier A1 at node 102. Thenodes 101 and 102 are driven by an output Vs of the periodic varyingvoltage source G1, which may be a sinusoidal voltage source, throughresistors R1 and R2, respectively. Resistor R1 has a resistance r1, andresistor R2 has a resistance r2. Output V1 of operational amplifier A1is fed back to the inverting input of operational amplifier A1 viafeedback resistor R3 having a resistance of r3. Because operationalamplifier A1 has a very high open loop gain, the two inputs ofoperational amplifier A1 are maintained at substantially the samevoltage potential, thereby causing the effective RC time constants forcapacitors C1 and C2 at nodes 101 and 102 to be substantially the same.The magnitude i3 of current I3 flowing through resistor R3 issubstantially equal to the magnitude i2 of current I2 flowing intocapacitor C2 multiplied by the factor (r2/r1-c1/c2), i.e.,i3=i2*(r2/r1-c1/c2). Current I3 flowing through resistor R3 will bein-phase or out-of-phase with current I1 flowing through resistor R1 andcurrent I2 flowing through resistor R2, depending on whether the ratioof the capacitances c1/c2 is less than or greater than the value r2/r1.More specifically, if c1/c2 is less than r2/r1, then (r2/r1-c1/c2) ispositive and the currents I2 and I3 will be in-phase but if c1/c2 isgreater than r2/r1, then (r2/r1-c1/c2) is negative and the currents I2and I3 will be out of phase. At steady state, the magnitude i2 ofcurrent I2 flowing into capacitor C2 is a function of time, andtherefore the magnitude i3 of current I3 flowing through resistor R3 ata fixed time of a cycle of current I3 is an accurate measure of thevalue (r2/r1-c1/c2). The voltage across resistor R3 is equal to i3*r3,and is equivalent to the difference of the voltage potential betweennode 101 (or node 102) and output V1 of operational amplifier A1. Thisvoltage can be measured by connecting node 101 (or node 102) to one ofthe two inputs of difference amplifier A2, and by connecting output V1of operational amplifier A1 to the other input of difference amplifierA2. It should be noted that the configuration of difference amplifierA2, as shown in FIG. 2 a, is described herein for purposes ofillustration, and that other suitable circuit configurations may beemployed. For example, alternative configurations of differenceamplifier A2 may include more than one operational amplifier. Output Vdof difference amplifier A2 is proportional to the magnitude i3 ofcurrent I3 flowing through resistor R3, and will be in-phase withcurrents I1 and I2 when c1/c2 is greater than r2/r1. It is noted thatthe phase of output Vd reverses if the inputs to difference amplifier A2are interchanged.

Output Vd of difference amplifier A2 is therefore proportional to asignal representing current I2 modulated by the value (c1/c2-r2/r1). Ifcurrent I2 is sinusoidal, then the change in the ratio of thecapacitances c1/c2 can be measured using a synchronous demodulator.Further, there is a linear relationship between output Vd at a fixedtime of a cycle of the output (e.g., at the peak of the cycle), theaverage absolute value of its positive and/or negative cycles, or thesignal extracted from output Vd using synchronous demodulation (ifoutput Vs of the voltage source G1 is sinusoidal), and the ratio of thecapacitances c1/c2. Accordingly, there is a linear relationship betweenoutput Vd at a fixed time of a cycle of the output, the average absolutevalue of its positive and/or negative cycles, or the signal extractedfrom the output using synchronous demodulation, and the capacitance c1if capacitor C2 has a fixed capacitance, or the reciprocal of thecapacitance c2 if capacitor C1 has a fixed capacitance, which isparticularly useful when measuring distances because the capacitancebetween two conductive objects, e.g., two plates, is inverselyproportional to the distance between them.

FIG. 2 b depicts circuitry 200 b employing capacitance measurementcircuitry 200 a 1-200 an to produce a linear response to changes in eachof the ratios of the capacitance of a capacitor/capacitive transducer tothe capacitance of one or more other different capacitors/capacitivetransducers, while keeping all of the capacitors/capacitive transducersat substantially identical voltage potentials at all times. Each of aplurality of capacitors/capacitive transducers C11-C1 n havingcapacitances of c11-c1 n, respectively, is compared to the capacitancec2 of capacitor/capacitive transducer C2 coupled between thenon-inverting input of an operational amplifier A0 and ground (see FIG.2 b). Each of the capacitance measurement circuitry 200 a 1-200 anoperates like the capacitance measurement circuitry 200 a (see FIG. 2a), with the exception that the voltage potentials at capacitors C11-C1n are compared to the level of the output of operational amplifier A0,which is configured as a voltage follower to produce substantially thesame voltage potential as that across capacitor C2. It is noted thatcapacitor C2 is driven by output Vs of the voltage source G1 throughresistor R2. Thus, there is a linear relationship between each ofoutputs Vd1-Vdn at a fixed time of a cycle of the output (e.g., at thepeak of the cycle), the average absolute value of its positive and/ornegative cycles, or the signal extracted from the output usingsynchronous demodulation (if output Vs of the voltage source G1 issinusoidal),. and the ratio of the capacitances c11/c2 through c1 n/c2,respectively. As a result, there is a linear relationship between eachof outputs Vd1-Vdn at a fixed time of a cycle of the output, the averageabsolute value of its positive and/or negative cycles, or the signalextracted from the output using synchronous demodulation, and thecapacitances c11-c1 n, respectively, if C2 has a fixed capacitance.

FIG. 3 a depicts a second illustrative embodiment of capacitancemeasurement circuitry 300 a, in accordance with the present invention.In the illustrated embodiment, the capacitance measurement circuitry 300a includes periodic varying current sources G1 and G2, a firstoperational amplifier A1, and a second operational amplifier A2configured as a difference amplifier. Each of two capacitors/capacitivetransducers C1 and C2 undergoing comparison, having capacitances c1 andc2, respectively, has one end connected to circuit ground, and anotherend connected to one of the differential inputs of operational amplifierA1. Capacitor C1 is connected to the inverting input of operationalamplifier Al at node 101, and capacitor C2 is connected to thenon-inverting input of operational amplifier A1 at node 102. The node102 is driven by output current I2 of the periodic varying currentsource G2. The node 101 is driven by output current I1 of the periodicvarying current source G1, in which I1 is equal to K times I2, K being aconstant greater than or equal to zero. Output V1 of operationalamplifier A1 is fed back to its inverting input via feedback resistor R1having a resistance r1. Because operational amplifier A1 has a very highopen loop gain, the two inputs of operational amplifier A1 aremaintained at substantially the same voltage potential. Therefore, themagnitude i3 of current I3 flowing through resistor R1 is substantiallyequal to the magnitude i2 of current I2 flowing into capacitor C2multiplied by the factor (K-c1/c2), i.e., i3=i2*(K-c1/c2). Current I3flowing through resistor R1 will be in-phase or out-of-phase withcurrents I1 and I2, depending on whether the ratio of the capacitancesc1/c2 is less than or greater than K. More specifically, if c1/c2 isless than K, then (K-c1/c2) is positive and currents I2 and I3 will bein-phase but if c1/c2 is greater than K, then (K-c1/c2) is negative andcurrents I2 and I3 will be out of phase. At steady state, the magnitudei2 of current I2 flowing into capacitor C2 is only a function of time,and therefore the magnitude i3 of current I3 flowing through resistor R1at a fixed time of a cycle of the current I3 is an accurate measure ofthe value (K-c1/c2). The voltage across resistor R1 is substantiallyequal to i3*r1, and is equivalent to the difference of the voltagepotential between node 101 (or node 102) and output V1 of operationalamplifier A1. The voltage across resistor R1 can be measured byconnecting node 101 (or node 102) to one of the two inputs of differenceamplifier A2, and by connecting output V1 of operational amplifier A1 tothe other input of difference amplifier A2. It should be noted that theconfiguration of difference amplifier A2, as shown in FIG. 3 a, isdescribed herein for purposes of illustration, and that other suitablecircuit configurations may be employed. For example, alternativeconfigurations of difference amplifier A2 may include more than oneoperational amplifier. Output Vd of difference amplifier A2 isproportional to the magnitude i3 of current I3 flowing through resistorR1, and will be in-phase with currents I1 and I2 when the ratio of thecapacitances c1/c2 is greater than K. It is noted that the phase ofoutput Vd reverses if the inputs to difference amplifier A2 areinterchanged. Output Vd of difference amplifier A2 is thereforeproportional to a signal representing current I2 modulated by the value(c1/c2-K). In the event the constant K equals 0, i.e., there is nocurrent source G1, output Vd of difference amplifier A2 is proportionalto a signal representing current I2 modulated by the value c1/c2. Ifcurrent I2 is sinusoidal, then the change in the ratio of thecapacitances c1/c2 can be measured using a synchronous demodulator forall values of K. Further, there is a linear relationship between outputVd at a fixed time of a cycle of the output (e.g., at the peak of thecycle), the average absolute value of its positive and/or negativecycles, or the signal extracted from the output using synchronousdemodulation (if output I2 of the current source G2 is sinusoidal), andthe ratio of the capacitances c1/c2. Thus, there is alinear-relationship between output Vd at a fixed time of a cycle of theoutput, the average absolute value of its positive and/or negativecycles, or the signal extracted from the output using synchronousdemodulation, and the capacitance c1 if capacitor C2 has a fixedcapacitance, or the reciprocal of the capacitance c2 if capacitor C1 hasa fixed capacitance, which is particularly useful when measuringdistances because the capacitance between two conductive objects, e.g.,two plates, is inversely proportional to the distance between them. Itshould be noted that if the current sources G1 and G2 have significantdc components in their outputs, then bypass resistors may be placedacross capacitors C1 and C2.

FIG. 3 b depicts circuitry 300 b employing capacitance measurementcircuitry 300 al-300 an to produce a linear response to changes in eachof the ratios of the capacitance of a capacitor/capacitive transducer tothe capacitance of one or more other different capacitors/capacitivetransducers, while keeping all of the capacitors/capacitive transducersat substantially identical voltage potentials at all times. Each of thecapacitors/capacitive transducers C11-C1 n having capacitances c11-c1 n,respectively, is compared to the capacitance c2 of capacitor/capacitivetransducer C2 coupled between the non-inverting input of operationalamplifier A0 and ground (see FIG. 3 b). Capacitors C11-C1 n are drivenby current sources G11-G1 n, respectively. Each of the respectiveoutputs I11-I1 n of the current sources G11-G1 n is equal to I2 times arespective constant K11-K1 n, in which each of the constants K11-K1 n isgreater than or equal to zero. Each of the capacitance measurementcircuitry 300 a 1-300 an operates like the capacitance measurementcircuitry 300 a (see FIG. 3 a), with the exception that the voltagepotentials at capacitors C11-C1 n are compared to the level of theoutput of operational amplifier A0, which is configured as a voltagefollower to produce substantially the same voltage potential as thatacross capacitor C2, which is driven by output I2 of current source G2.Therefore, there is a linear relationship between each of outputsVd1-Vdn at a fixed time of a cycle of the output (e.g., at the peak ofthe cycle), the average absolute value of its positive and/or negativecycles, or the signal extracted from the output using synchronousdemodulation (if output I2 of the current source G2 is sinusoidal), andthe ratio of the capacitances cll/c2 through c1 n/c2, respectively. As aresult, there is a linear relationship between each of outputs Vd1-Vdnat a fixed time of a cycle of the output, the average absolute value ofits positive and/or negative cycles, or the signal extracted from theoutput using synchronous demodulation, and the capacitances c11-c1 n,respectively, if C2 has a fixed capacitance.

FIG. 4 a depicts a third illustrative embodiment of capacitancemeasurement circuitry 400 a, in accordance with the present invention.In the illustrated embodiment, the capacitance measurement circuitry 400a includes a periodic varying voltage source G1, a first operationalamplifier A0, a second operational amplifier A1, and a third operationalamplifier A2 configured as a difference amplifier. Each of twocapacitors/capacitive transducers C1 and C2 undergoing comparison,having capacitances c1 and c2, respectively, has one end connected tocircuit ground, and another end connected to the inverting input ofoperational amplifier A0 or operational amplifier A1. Capacitor C1 isconnected to the inverting input of operational amplifier A1 at node101, and capacitor C2 is connected to the inverting input of operationalamplifier A0 at node 102. A first feedback resistor R1 having resistancer1 is connected between the output of operational amplifier A1 and itsinverting input. Similarly, a second feedback resistor R2 havingresistance r2 is connected between the output of operational amplifierA0 and its inverting input. The non-inverting inputs of operationalamplifiers A1 and A0 are both driven by output Vs of the periodicvarying voltage source G1, which may be a sinusoidal voltage source. Dueto the high open loop gain of operational amplifiers A1 and A0,capacitors C1 and C2 are maintained at substantially the same voltagepotential as output Vs of the voltage source G1 at all times. V0 is theoutput of the operational amplifier A0 and V1 is the output of theoperational amplifier A1. (Vl-Vs) is equal to the time derivative of Vsmultiplied by the value (r1*c1), i.e., (V1-Vs)=r1*c1*dVs/dt, and isin-phase with current I1 flowing through resistor R1 into capacitor C1.(V0-Vs) is equal to the time derivative of Vs multiplied by the value(r2*c2), i.e., (V0-Vs)=r2*c2*dVs/dt, and is in-phase with current I2flowing through resistor R2 into capacitor C2. When output V1 ofoperational amplifier A1 is provided to the non-inverting input ofdifference amplifier A2, while output V0 of operational amplifier A0 isprovided to the inverting input of difference amplifier A2, output Vd ofdifference amplifier A2 is proportional to a signal representing thetime derivative of Vs modulated by the value (r1*c1-r2*c2), and isin-phase or out-of-phase with current flowing through resistors R1 andR2, depending on whether (r1*c1-r2*c2) is greater than or less than zero(the phase of output Vd reverses if the inputs to difference amplifierA2 are interchanged). It should be noted that the configuration ofdifference amplifier A2, as shown in FIG. 4 a, is described herein forpurposes of illustration, and that other suitable circuit configurationsmay be employed. For example, alternative configurations of differenceamplifier A2 may include more than one operational amplifier. If voltageVs is sinusoidal, then the change in the value of (r1*c1-r2*c2) can bemeasured using a synchronous demodulator. At steady state, the timederivative of Vs is only a function of time, and therefore there is alinear relationship between output Vd at a fixed time of a cycle of theoutput (e.g., at the peak of the cycle), the average absolute value ofits positive and/or negative cycles, or the signal extracted from theoutput using synchronous demodulation (if output Vs of the voltagesource G1 is sinusoidal), and the value (r1*c1-r2*c2). As a result,there is a linear relationship between output Vd at a fixed time of acycle of the output, the average absolute value of its positive and/ornegative cycles, or the signal extracted from the output usingsynchronous demodulation, and the capacitance c1 if capacitor C2 has afixed capacitance, or the capacitance c2 if capacitor C1 has a fixedcapacitance.

FIG. 4 b depicts circuitry 400 b employing capacitance measurementcircuitry 400 a 1-400 an to produce a linear response to changes in eachof the differences between the capacitance of a capacitor/capacitivetransducer multiplied by a first constant factor, and the capacitance ofone or more other different capacitors/capacitive transducers after eachis multiplied by a respective second constant factor, while keeping allof the capacitors/capacitive transducers at substantially identicalvoltage potentials at all times. Each of capacitors/capacitivetransducers C11-C1 n having capacitances of c11-c1 n, respectively, iscompared to the capacitance c2 of capacitor/capacitive transducer C2.Each of the capacitance measurement circuitry 400 a 1-400 an inconjunction with operational amplifier A0 feedback resistor R2, andcapacitor C2, operates like the capacitance measurement circuitry 400 a(see FIG. 4 a). There is therefore a linear relationship between each ofoutputs Vd1-Vdn at a fixed time of a cycle of the output (e.g., at thepeak of the cycle), the average absolute value of its positive and/ornegative cycles, or the signal extracted from the output usingsynchronous demodulation (if output Vs of the voltage source G1 issinusoidal), and its respective value (r1 n*c1 n-r2*c2), in which “r1 n”is the resistance of the respective feedback resistor R1 n associatedwith the respective operational amplifier A1 n. Accordingly, there is alinear relationship between each of outputs Vd 1-Vdn at a fixed time ofa cycle of the output, the average absolute value of its positive and/ornegative cycles, or the signal extracted from the output usingsynchronous demodulation, and the capacitances c11-c1 n, respectively,if capacitor C2 has a fixed capacitance.

By switching capacitors/capacitive transducers in and out for subsequentmeasurement, each embodiment of the presently disclosed capacitancemeasurement circuitry can sequentially produce linear responses tochanges in the capacitance or the reciprocal of the capacitance of alarge number of capacitors/capacitive transducers, or compare thecapacitances of a large number of capacitors/capacitive transducers tothe capacitances of a large number of sets of capacitors/capacitivetransducers. It is noted that any suitable type of capacitive transducermay be employed in each embodiment of the capacitance measurementcircuitry described above, including but not limited to any suitabletype of capacitive transducer for sensing force, pressure, strain,acceleration, sound, mechanical displacement, fluid flow, etc. It isfurther noted that each embodiment of the capacitance measurementcircuitry described above may employ any suitable type of double-endedpower supply, or single-ended power supply, if an appropriate circuitground reference can be provided e.g., by a voltage splitter circuit.

FIGS. 5 a-5 d depict illustrative embodiments of a front surface of atouchless switch, in accordance with the present invention. The frontsurface of the presently disclosed touchless switch can take the form ofany suitable type of container such as containers 500 a-500 c depictedin FIGS. 5 a-5 c, respectively. As shown in FIGS. 5 a-5 c, each of thecontainers 500 a-500 c includes a base portion such as base portions 502a-502 c of FIGS. 5 a-5 c, respectively, and a brim portion such as brimportions 504 a-504 c of FIGS. 5 a-5 c, respectively. Alternatively, thefront surface of the switch can be flat like a front surface 500 d (seeFIG. 5 d), or any other suitable surface configuration. The presentlydisclosed touchless switch includes two sensor electrodes, specifically,a center electrode and an outer electrode, which are disposed on orbehind the front surface of the switch and are maintained atsubstantially the same voltage potential. The center electrode is spacedfrom and at least partly surrounded by the outer electrode. The centerand outer electrodes can be of any suitable shape, form, or size, andneed not be a solid piece, e.g., an electrode may be a wire mesh. FIGS.6 a-6 d depict illustrative arrangements and positions of the center andouter electrodes relative to each other, and relative to the frontsurfaces 500 a-500 d of FIGS. 5 a-5 d, respectively. As shown in FIGS. 6a-6 c, when the front surface is in the form of a container, the centerelectrode is disposed near the base of the container, and the outerelectrode is disposed near the brim of the container.

The presence of the tip of a human finger or a finger-like object nearor proximate to the center electrode of the touchless switch can bedetected using an embodiment of the presently disclosed capacitancemeasurement circuitry. When detecting the presence of the human fingeror finger-like object, the capacitance measurement circuitry comparesthe capacitances of the capacitors formed between the two sensorelectrodes (i.e., the center electrode and the outer electrode) and thehuman finger or finger-like object, thereby substantially canceling outextraneous common-mode disturbances, for example, the capacitancebetween the rest of the human body and the sensor electrodes, thecapacitance between the human body and ground, environmental changes,electrical noise, etc. Such extraneous common-mode disturbances tend toaffect both sensor electrodes equally due to their close proximity toone another. Additionally, the outer electrode can be positioned in thetouchless switch so that the tip of the human finger or finger-likeobject is required to go past a specified boundary before actuating theswitch. In one embodiment, this is accomplished by positioning the outerelectrode a specified distance in front of the center electrode, andconfiguring the spacing between the two sensor electrodes and theirrelative surface areas so that when an object is near the electrodes,but more than a specified distance away from the center electrode, theratio of the capacitance between the object and the center electrode tothe capacitance between the object and the outer electrode, or thedifference between the capacitance between the object and the centerelectrode multiplied by a first constant and the capacitance between theobject and the outer electrode multiplied by a second constant, is lessthan a preset threshold. It is understood that the capacitance ratio anddifference measurements are performed by an embodiment of theabove-described capacitance measurement circuitry, and are facilitatedby the fixed geometrical shape, size, and relative position of the twosensor electrodes. In one embodiment, the touchless switch is actuatedwhen the measured capacitance ratio or difference exceeds the presetthreshold.

Accordingly, the touchless switch is not actuated by a human finger orfinger-like object until the finger passes through the specifiedboundary. In the event the front surface of the touchless switch has theform of the surface of a container (see, e.g., FIGS. 5 a-5 c), thespecified boundary coincides with an imaginary plane defined by the brimof the container. As the tip of a human finger or finger-like objectmoves toward the center electrode and breaks the plane of the specifiedboundary, the capacitance associated with the center electrode increasesmore rapidly than the capacitance associated with the outer electrode.The touchless switch is actuated when the ratio of the capacitancesassociated with the center electrode and the outer electrode, or thedifference between the capacitance associated with the center electrodemultiplied by a first constant and the capacitance associated with theouter electrode multiplied by a second constant, exceeds the presetthreshold.

FIG. 7 depicts an illustrative embodiment of a touchless switch 700, inaccordance with the present invention. In the illustrated embodiment,the touchless switch 700 includes a front surface 702 in the form of acontainer, a center electrode 704, an outer electrode 706, and a guardelectrode 708 surrounding the back and the sides of the two sensorelectrodes 704, 706. All of the electrodes 704, 706, 708 are maintainedat substantially the same voltage potential. As a result, the two sensorelectrodes 704, 706 are operative to form electric fields only betweenthe sensor electrodes and objects disposed in front of the switch, i.e.,above the switch 700, as depicted in FIG. 7. Leads extending from thetwo sensor electrodes to the capacitance measurement circuitry may alsobe guarded using a twin-axial cable or two coaxial cables, in which theouter conductors of the cables are employed as guard shields andmaintained at substantially the same voltage potential as the insidecable conductors connected to the sensor electrodes.

FIG. 8 a depicts a first illustrative circuit implementation 800 a of atouchless switch, in accordance with the present invention. As shown inFIG. 8 a, the circuit implementation 800 a comprises a center electrodeE1, an outer electrode E2, a guard electrode E3, a startup delay section203, a switching decision section 205, a switching output section 207,and capacitance measurement circuitry 802 a, which includes a periodicvarying voltage source G1, operational amplifiers A0 and A1, resistorsR1-R3, and an operational amplifier A2 configured as a differenceamplifier. The center electrode El is connected to the inverting inputof operational amplifier A1 at node 201, and the outer electrode E2 isconnected at node 202 to the non-inverting input of operationalamplifier A0, which is configured as a voltage follower to provide thevoltage potential of outer electrode E2 to the non-inverting input ofoperational amplifier A1. The nodes 201 and 202 are both driven by anoutput Vs of the periodic varying voltage source G1. Electrodes E1 andE2 correspond to capacitors C1 and C2 of FIG. 2 a, respectively. Guardelectrode E3 is connected to the output of operational amplifier A0, andtherefore the voltage potential of guard electrode E3 is substantiallyequal to the voltage potentials of sensor electrodes E1 and E2. Guardelectrode E3 may be configured to surround the back and the sides ofsensor electrodes E1 and E2 so that capacitances can only be formedbetween conductive objects disposed in front of the touchless switch andsensor electrodes E1 and E2. It is noted that operational amplifiers A1and A2 of FIG. 8 a are like operational amplifiers A1 and A2 of FIG. 2a, respectively, resistor R2 of FIG. 8 a is like resistor R2 of FIG. 2a, resistors R1 and R3 of FIG. 8 a are like resistors R1 and R3 of FIG.2 a, respectively, and the periodic varying voltage source G1 of FIG. 8a is like the periodic varying voltage source G1 of FIG. 2 a. Thus,there is a linear relationship between output Vd of difference amplifierA2 at a fixed time of a cycle of the output (e.g., at the peak of thecycle), or the average absolute value of its positive and/or negativecycles, or the signal extracted from the output using synchronousdemodulation (if output Vs of voltage source G1 is sinusoidal), and theratio of the capacitance associated with center electrode E1 to thecapacitance associated with outer electrode E2. Difference amplifier A2provides output Vd to the switching decision section 205, whichdetermines whether to actuate the touchless switch based on signal Vd.For example, the switching decision section 205 can base its decision onthe phase, the amplitude, an average, and/or any other suitable propertyof signal Vd. Alternatively, the switching decision section 205 canrequire a specified number of consecutive detections of the requiredphase and/or amplitude of signal Vd, or the satisfaction of certaincriteria, before deciding to actuate the touchless switch. If voltage Vsis sinusoidal, then a synchronous demodulator can be included in theswitching decision section 205 so that the change in the ratio of thecapacitance associated with center electrode E1 to the capacitanceassociated with outer electrode E2 can be obtained with a high degree ofaccuracy, even at a high noise level. It should be noted that theswitching decision section 205 may require one or more signals inaddition to signal Vd to determine whether or not to actuate the switch.For example, the switching decision section 205 may require a referencesignal to determine the phase of signal Vd. The switching decisionsection 205 provides a logic signal 206 representing its decision to theswitching output section 207, which implements the required switchingaction. It is noted that the switching output section 207 may beimplemented using any suitable number of logical outputs (normally highor low), solid state switch outputs, and/or dry contact outputs(normally open or closed) in any suitable switching mode, including butnot limited to pulse mode, momentarily mode, toggle mode, etc. Theswitching output section 207 can also be configured to produce audioand/or visual outputs to indicate the status of the touchless switch.Because the capacitance measurement circuitry 802 a takes several cyclesof output Vs of the voltage source G1 to stabilize, the startup delaysection 203 outputs a startup signal 204 to the switching decisionsection 205 during the startup period to prevent the switching decisionsection 205 from inadvertently actuating the switch. When sensorelectrodes E1 and E2 are disposed at a distance away from the inputs ofoperational amplifiers A0 and A1, the leads from center electrode E1 andouter electrode E2 may be guarded using a twin-axial cable or twocoaxial cables of equal length, using the outer conductors as the guardshields connected to guard electrode E3 and maintained at substantiallythe same voltage potential as the inside conductors connected torespective sensor electrodes E1 and E2, so that no stray capacitance isintroduced and any other unwanted effects introduced by the leads aresubstantially cancelled out.

FIG. 8 b depicts a first illustrative circuit implementation 800 b of aset of touchless switches, including a periodic varying voltage sourceG1, an operational amplifier A0, capacitance measurement circuitry 802 a1-802 an, the startup delay section 203, the switching decision section205, and the switching output section 207. It is noted that each of thecapacitance measurement circuitry 802 a 1-802 an in conjunction withoperational amplifier A0 is like the capacitance measurement circuitry802 a (see FIG. 8 a), and corresponds to a respective touchless switchin the set of touchless switches. Specifically, electrode E2 coupled tothe non-inverting input of operational amplifier A0 corresponds to acommon outer electrode of the set of touchless switches, and electrodeE3 coupled to the output of operational amplifier A0 corresponds to acommon guard electrode of the set of touchless switches. Each ofelectrodes E11-E1 n corresponds to a center electrode of a respectivetouchless switch. It is noted that operational amplifier A0 of FIG. 8 bis like operational amplifier A0 of FIG. 2 b, operational amplifiersA11-A1 n of FIG. 8 b are like operational amplifiers A11-A1 n of FIG. 2b, respectively, difference amplifiers A2 l-A2 n of FIG. 8 b are likedifference amplifiers A2 l-A2 n of FIG. 2 b, respectively, resistor R2of FIG. 8 b is like resistor R2 of FIG. 2 b, resistors R1 l-R1 n of FIG.8 b are like resistors R1 l-R1 n of FIG. 2 b, respectively, resistors R3l-R3 n of FIG. 8 b are like resistors R3 l-R3 n of FIG. 2 b,respectively, and the periodic varying voltage source G1 of FIG. 8 b islike the periodic varying voltage source G1 of FIG. 2 b. Differenceamplifiers A2 l-A2 n provide output signals Vd1-Vdn, respectively, tothe switching decision section 205, which determines when to actuateeach switch based on the respective signals Vd1-Vdn. The switchingdecision section 205 provides logic signals 206 representing itsrespective decisions to the switching output section 207, whichimplements the required switching action for each switch. It is notedthat the switching output section 207 may be implemented using anysuitable number of logical outputs (normally high or low), solid stateswitch outputs, and/or dry contact outputs (normally open or closed) inany suitable switching mode, including but not limited to pulse mode,momentarily mode, toggle mode, etc., for each switch. The switchingoutput section 207 can also be configured to produce audio and/or visualoutputs to indicate the status of each switch. The startup delay section203 of FIG. 8 b is like the corresponding section 203 described abovewith reference to FIG. 8 a, and each switch of circuit implementation800 b (see FIG. 8 b) basically operates like the switch of circuitimplementation 800 a (see FIG. 8 a).

FIG. 9 a depicts a second illustrative circuit implementation 900 a of atouchless switch, in accordance with the present invention. As shown inFIG. 9 a, the circuit implementation 900 a comprises a center electrodeE1, an outer electrode E2, a guard electrode E3, a startup delay section203, a switching decision section 205, a switching output section 207,and capacitance measurement circuitry 902 a, which includes periodicvarying current sources G1 and G2, operational amplifiers A0 and A1,resistor R1, and an operational amplifier A2 configured as a differenceamplifier. The center electrode E1 is connected to the inverting inputof operational amplifier A1 at node 201, and outer electrode E2 isconnected at node 202 to the non-inverting input of operationalamplifier A0, which is configured as a voltage follower to provide thevoltage potential of outer electrode E2 to the non-inverting input ofoperational amplifier A1. The node 201 is driven by an output current I1of the periodic varying current source G1 and node 202 is driven by anoutput current I2 of the periodic varying current source G2. ElectrodesE1 and E2 correspond to capacitors C1 and C2 of FIG. 3 a, respectively.Guard electrode E3 is connected to the output of operational amplifierA0, and therefore the voltage potential of guard electrode E3 issubstantially the same as the voltage potentials of sensor electrodes E1and E2. Guard electrode E3 may be configured to surround the back andthe sides of sensor electrodes E1 and E2 so that capacitances can beformed only between conductive objects disposed in front of thetouchless switch and the sensor electrodes E1 and E2. It is noted thatoperational amplifiers A1 and A2 of FIG. 9 a are like operationalamplifiers Al and A2 of FIG. 3 a, respectively, resistor R1 of FIG. 9 ais like resistor R1 of FIG. 3 a, and the periodic varying currentsources G1 and G2 of FIG. 9 a are like the periodic varying currentsources G1 and G2 of FIG. 3 a, respectively. Thus, there is a linearrelationship between output Vd of difference amplifier A2 at a fixedtime of a cycle of the output (e.g., at the peak of the output), or theaverage absolute value of its positive and/or negative cycles, or thesignal extracted from the output using synchronous demodulation (ifoutput I2 of current source G2 is sinusoidal), and the ratio of thecapacitance associated with center electrode E1 to the capacitanceassociated with outer electrode E2. Difference amplifier A2 providesoutput signal Vd to the switching decision section 205, which determineswhether to actuate the touchless switch based on signal Vd. For example,the switching decision section 205 can base its decision on the phase,the amplitude, an average, and/or any other suitable property of signalVd. Alternatively, the switching decision section 205 can require aspecified number of consecutive detections of the required phase and/oramplitude of signal Vd, or the satisfaction of certain criteria, beforedeciding to actuate the switch. If current I2 is sinusoidal, then asynchronous demodulator can be included in the switching decisionsection 205 so that the change in the ratio of the capacitanceassociated with center electrode E1 to the capacitance associated withouter electrode E2 can be obtained with a high degree of accuracy, evenat a high noise level. It should be noted that the switching decisionsection 205 may require one or more signals in addition to signal Vd todetermine whether or not to actuate the switch. For example, theswitching decision section 205 may require a reference signal todetermine the phase of signal Vd. The switching decision section 205provides a logic signal 206 representing its decision to the switchingoutput section 207, which implements the required switching action. Itis noted that the switching output section 207 may be implemented usingany suitable number of logical outputs (normally high or low), solidstate switch outputs, and/or dry contact outputs (normally open orclosed) in any switching mode, including but not limited to pulse mode,momentarily mode, toggle mode, etc. The switching output section 207 canalso be configured to produce audio and/or visual outputs to indicatethe status of the switch. Because the capacitance measuring circuitry902 a takes several cycles of the output I2 of the periodic varyingcurrent source G2 to stabilize, a startup delay section 203 provides astartup signal 204 to the switching decision section 205 during thestartup period to prevent it from inadvertently actuating the switch.When sensor electrodes E1 and E2 are disposed at a distance away fromthe inputs of operational amplifiers A0 and A1, the leads from sensorelectrodes E1 and E2 may be guarded using a twin-axial cable or twocoaxial cables of equal length, with the outer conductors employed asguard shields connected to guard electrode E3 and maintained atsubstantially the same voltage potential as the inside conductorsconnected to respective sensor electrodes E1 and E2, so that no straycapacitance is introduced and any other unwanted effects introduced bythe leads are substantially cancelled out.

FIG. 9 b depicts a second illustrative circuit implementation 900 b of aset of touchless switches, including periodic varying current sourcesG11-G1 n, a periodic varying current source G2, an operational amplifierA0, capacitance measurement circuitry 902 a 1-902 an, the startup delaysection 203, the switching decision section 205, and the switchingoutput section 207. It is noted that each of the capacitance measurementcircuitry 902 a 1-902 an, in conjunction with operational amplifier A0,is like the capacitance measurement circuitry 902 a (see FIG. 9 a), andcorresponds to a respective touchless switch in the set of touchlessswitches. Specifically, electrode E2 coupled to the non-inverting inputof operational amplifier A0 corresponds to a common outer electrode ofthe set of touchless switches, and electrode E3 coupled to the output ofoperational amplifier A0 corresponds to a common guard electrode of theset of touchless switches. Each of electrodes E11-E1 n corresponds to acenter electrode of a respective touchless switch. Further, operationalamplifier A0 of FIG. 9 b is like operational amplifier A0 of FIG. 3 b,operational amplifiers A11-A1 n of FIG. 9 b are like operationalamplifiers A11-A1 n of FIG. 3 b, respectively, difference amplifiersA21-A2 n of FIG. 9 b are like difference amplifiers A21-A2 n of FIG. 3b, respectively, resistors R11-R1 n of FIG. 9 b are like resistorsR11-R1 n of FIG. 3 b, respectively, the periodic varying current sourceG2 of FIG. 9 b is like the periodic varying current source G2 of FIG. 3b, and the periodic varying current sources G11-G1 n of FIG. 9 b arelike the periodic varying current sources G11-G1 n of FIG. 3 b.Difference amplifiers A21-A2 n provide output signals Vd1-Vdn,respectively, to the switching decision section 205, which determineswhen to actuate each switch based on the respective signals Vd1-Vdn. Theswitching decision section 205 provides logic signals 206 representingits respective decisions to the switching output section 207, whichimplements the required switching action for each switch. It is notedthat the switching output section 207 may be implemented using anysuitable number of logical outputs (normally high or low), solid stateswitch outputs, and/or dry contact outputs (normally open or closed) inany suitable switching mode, including but not limited to pulse mode,momentarily mode, toggle mode, etc., for each switch. The switchingoutput section 207 can also be configured to produce audio and/or visualoutputs to indicate the status of each switch. The startup delay section203 of FIG. 9 b is like the startup delay section 203 of FIG. 9 a, andeach switch of circuit implementation 900 b (see FIG. 9 b) basicallyoperates like the switch of circuit implementation 900 a (see FIG. 9 a).

FIG. 10 a depicts a third illustrative circuit implementation 1000 a ofa touchless switch, in accordance with the present invention. As shownin FIG. 10 a, the circuit implementation 1000 a comprises a centerelectrode E1, an outer electrode E2, a guard electrode E3, a startupdelay section 203, a switching decision section 205, a switching outputsection 207, and capacitance measurement circuitry 1002 a, whichincludes a periodic varying voltage source G1, operational amplifiers A0and A1, resistors R1 and R2, and an operational amplifier A2 configuredas a difference amplifier. The non-inverting inputs of operationalamplifier A0 and A1 are both driven by an output Vs of the periodicvarying voltage source G1. The center electrode E1 is connected to theinverting input of operational amplifier A1 at node 201, and outerelectrode E2 is connected at node 202 to the inverting input ofoperational amplifier A0. It is noted that sensor electrodes E1 and E2correspond to capacitors C1 and C2 of FIG. 4 a, respectively. Guardelectrode E3 is connected to the output of the periodic varying voltagesource G1, and is therefore substantially the same as the voltagepotential of the two sensor electrodes E1 and E2. Guard electrode E3 maybe configured to surround the back and the sides of the sensorelectrodes E1 and E2 so that capacitances can be formed only betweenconductive objects disposed in front of the touchless switch and thesensor electrodes E1 and E2. It is noted that operational amplifiers A0and A1 of FIG. 10 a are like operational amplifiers A0 and A1 of FIG. 4a, respectively, resistor R1 of FIG. 10 a is like resistor R1 of FIG. 4a, resistor R2 of FIG. 10 a is like resistor R2 of FIG. 4 a, and theperiodic varying voltage source G1 of FIG. 10 a is like the periodicvarying voltage source G1 of FIG. 4 a. Thus, there is a linearrelationship between output Vd of difference amplifier A2 at a fixedtime of a cycle of the output (e.g., at the peak of the cycle), or theaverage absolute value of its positive and/or negative cycles, or thesignal extracted from the output using synchronous demodulation (ifoutput Vs of the voltage source G1 is sinusoidal) and the value(r1*c1-r2*c2), in which r1 and r2 are the respective resistances ofresistors R1 and R2 and c1 and c2 are the respective capacitancesassociated with sensor electrodes E1 and E2. The difference amplifier A2provides output signal Vd to the switching decision section 205, whichdetermines whether to actuate the switch based on signal Vd. Forexample, the switching decision section 205 can base its decision on thephase, the amplitude, an average, and/or any other suitable property ofsignal Vd. Alternatively, the switching decision section 205 can requirea specified number of consecutive detections of the required phaseand/or amplitude of signal Vd, or the satisfaction of certain criteria,before deciding to actuate the switch. If voltage Vs is sinusoidal, thena synchronous demodulator can be included in the switching decisionsection 205 so that the change in the value of (r1*c1-r2*c2) can beobtained with a high degree of accuracy, even at a high noise level. Itshould be noted that the switching decision section 205 may require oneor more signals in addition to signal Vd to determine whether or not toactuate the switch. For example, the switching decision section 205 mayrequire a reference signal to determine the phase of output signal Vd.The switching decision section 205 provides a logic signal 206representing its decision to the switching output section 207, whichimplements the required switching action. It is noted that the switchingoutput section 207 may be implemented using any suitable number oflogical outputs (normally high or low), solid state switch outputs,and/or dry contact outputs (normally open or closed) in any switchingmode, including but not limited to pulse mode, momentarily mode, togglemode, etc. The switching output section 207 can also be configured toproduce audio and/or visual outputs to indicate the status of theswitch. Because the capacitance measuring circuit 1002 a takes severalcycles of output Vs of the periodic varying voltage source G1 tostabilize, a startup delay section 203 provides a startup signal 204 tothe switching decision section 205 during the startup period to preventit from inadvertently actuating the switch. When sensor electrodes E1and E2 are disposed at a distance away from the inputs of operationalamplifiers A0 and A1, the leads from sensor electrodes E1 and E2 may beguarded using a twin-axial cable or two coaxial cables of equal length,with the outer conductors employed as guard shields connected to theguard electrode E3 and maintained at substantially the same voltagepotential as the inside conductors connected to respective sensorelectrodes E1 and E2, so that no stray capacitance is introduced and anyother unwanted effects introduced by the leads are substantiallycancelled out.

FIG. 10 b depicts a third illustrative circuit implementation 1000 b ofa set of touchless switches, including a periodic varying voltage sourceG1, an operational amplifier A0, capacitance measurement circuitry 1002a 1-1002 an, the startup delay section 203, the switching decisionsection 205, and the switching output section 207. It is noted that eachof the capacitance measurement circuitry 1002 a 1-1002 an, inconjunction with operational amplifier A0, is like the capacitancemeasurement circuitry 1002 a (see FIG. 10 a), and corresponds to arespective touchless switch in the set of touchless switches.Specifically, electrode E2 coupled to the inverting input of operationalamplifier A0 corresponds to a common outer electrode of the set oftouchless switches, and electrode E3 coupled to the output Vs of voltagesource G1 corresponds to a common guard electrode of the set oftouchless switches. Each of electrodes E11-E1 n corresponds to a centerelectrode of a respective touchless switch. Further, operationalamplifier A0 of FIG. 10 b is like operational amplifier A0 of FIG. 4 b,operational amplifiers A11-A1 n of FIG. 10 b are like operationalamplifiers A11-A1 n of FIG. 4 b, respectively, difference amplifiersA21-A2 n of FIG. 10 b are like difference amplifiers A21-A2 n of FIG. 4b, respectively, resistor R2 of FIG. 10 b is like resistor R2 of FIG. 4b, resistors R11-R1 n of FIG. 10 b are like resistors R11-R1 n of FIG. 4b, respectively, and the periodic varying voltage source G1 of FIG. 10 bis like the periodic varying voltage source G1 of FIG. 4 b. Thedifference amplifiers A21-A2 n provide output signals Vd1-Vdn,respectively, to the switching decision section 205, which determineswhen to actuate each switch based on the respective signals Vd1-Vdn. Theswitching decision section 205 provides logic signals 206 representingits respective decisions to the switching output section 207, whichimplements the required switching action for each switch. It is notedthat the switching output section 207 may be implemented using anysuitable number of logical outputs (normally high or low), solid stateswitch outputs, and/or dry contact outputs (normally open or closed) inany suitable switching mode, including but not limited to pulse mode,momentarily mode, toggle mode, etc., for each switch. The switchingoutput section 207 can also be configured to produce audio and/or visualoutputs to indicate the status of each switch. The startup delay section203 of FIG. 10 b is like the startup delay section 203 of FIG. 10 a, andeach switch of circuit implementation 1000 b (see FIG. 10 b) operateslike the switch of circuit implementation 1000 a (see FIG. 10 a).

Having described the above illustrative embodiments, other alternativeembodiments or variations may be made. For example, each of thepresently disclosed circuit implementations of touchless switches can bescaled up to detect the proximity of a larger human appendage or otherconductive object, e.g., the palm of a human hand. The aboveillustrative embodiments can also be adapted to detect the position ormovement of a human appendage or conductive object by measuring thecapacitances between the object and each of an array of sensorelectrodes, using one of the capacitance measuring techniques describedabove and analyzing the results using suitable electronic circuitry or asuitably programmed computer.

It is noted that the outer electrode of the touchless switch may bepositioned in front of, behind, or at any other suitable positionrelative to the center electrode or set of center electrodes, dependingupon the specific application. Also, as discussed above, the electrodescan be of any suitable shape, form, or size, and need not be a solidpiece. For example, in an application for a proximity sensor in whichthe sensor is used in an outdoor environment, the proximity sensor canhave an outer electrode in the form of an insulated conductive meshdisposed on the outer surface of the proximity sensor, and a centerelectrode placed behind the outer electrode on the inner side of thesurface of the proximity sensor. In such an application, the presence ofa human appendage or other conductive object near or proximate to theproximity sensor in front of the center electrode causes the capacitanceassociated with the center electrode to increase more than thecapacitance associated with the outer electrode. The condition in whichthe relative changes in the capacitance associated with the centerelectrode and the capacitance associated with the outer electrode exceeda preset threshold can then be detected using one of the above-describedembodiments of the capacitance measurement circuitry. When water ormoisture is deposited on the outer surface of the proximity sensor, itis essentially deposited on the insulated conductive mesh of the outerelectrode. As a result, the water or moisture has a greater effect onthe capacitance associated with the outer electrode than the capacitanceassociated with the center electrode for the same amount of surface areadue to fact that it is much closer to the outer electrode, beingseparated by just the thickness of the insulation of the outerelectrode. Since the outer electrode is in the form of a mesh, thesurface area of the outer electrode in contact with the water ormoisture is usually much smaller than the area projected onto thesurface of the center electrode by the water or moisture. By properlydesigning the relative sizes of the surface areas and the spacingbetween the two electrodes of the proximity sensor, the relative changesin the capacitance associated with the two electrodes due to thepresence of water or moisture on the surface of the proximity sensor canbe made never to exceed the preset threshold, thereby enabling theproximity sensor to operate outdoors under inclement weather conditions.It is possible to have the insulated conductive mesh of the outerelectrode reside in grooved or recessed areas of the surface of theproximity sensor so that the insulated conductive mesh can come incontact with water or moisture deposited on the surface of the proximitysensor, but not a human appendage or other conductive object near theproximity sensor. In this way, the relative changes in the capacitanceassociated with the two electrodes due to a human appendage orconductive object touching the surface of the proximity sensor can bemade to exceed the preset threshold, while the mere presence of water ormoisture on the surface of the proximity sensor does not cause therelative changes in capacitance to exceed the preset threshold.

In addition, while the present invention may be embodied using hardwarecomponents, it is appreciated that one or more functions necessary toimplement the invention may alternatively be embodied in whole or inpart using hardware or software or some combination thereof usingmicro-controllers, microprocessors, digital signal processors,programmable logic arrays, or any other suitable hardware and/orsoftware.

It will be appreciated by those of ordinary skill in the art thatfurther modifications to and variations of the above-described linearcapacitance measurement and touchless switch may be made withoutdeparting from the inventive concepts disclosed herein. Accordingly, theinvention should not be viewed as limited except as by the scope andspirit of the appended claims.

1. A capacitive sensing apparatus for generating at least one logicalsignal in response to proximity of a conductive object, the capacitivesensing apparatus comprising: at least one first sensor electrode, afirst capacitance being induced by the conductive object in proximitywith the at least one first sensor electrode; a second sensor electrodedisposed near the at least one first sensor electrode, a secondcapacitance being induced by the conductive object in proximity with thesecond sensor electrode; and an apparatus configured to maintain the atleast one first sensor electrode and the second sensor electrode atsubstantially equal voltage potentials with respect to a common ground,and to measure relative changes in the first capacitance with respect tothe second capacitance, wherein the at least one logical signal isindicative of a status condition in which the relative changes in thefirst and second capacitances exceed at least one preset threshold. 2.The capacitive sensing apparatus of claim 1 wherein the at least onefirst sensor electrode and the second sensor electrode are each disposedon one of two sides of a surface, wherein the surface contains at leastone depressed area, each depressed area in a form of a container havinga base and a brim, wherein the at least one first sensor electrode isdisposed near the base of a corresponding container of the at least onedepressed area, and wherein the second sensor electrode is disposed nearthe brim of each container.
 3. The capacitive sensing apparatus of claim1 further comprising a guard electrode, wherein the guard electrode iselectrically isolated from the at least one first sensor electrode andthe second sensor electrode, wherein the guard electrode is maintainedat substantially the same voltage potential as the at least one firstsensor electrode and the second sensor electrode, and wherein the guardelectrode is configured to at least partly surround the at least onefirst sensor electrode and the second sensor electrode.
 4. Thecapacitive sensing apparatus of claim 1 wherein the second sensorelectrode is a conductive mesh and is disposed in front of the at leastone first sensor electrode.